Bit error rate reduction in chaotic communications

ABSTRACT

A system for chaotic sequence spread spectrum communications includes a transmitter ( 402 ) for transmitting information symbols using a chaotic sequence of chips generated at the transmitter, the information symbols having a duration of transmission based on a threshold symbol energy value and the chips. The system also includes a receiver ( 404 ) for extracting the information symbols from the transmitted signal using a chaotic sequence of chips generated at the receiver and the threshold symbol energy value. In the system, the chips generated at the transmitter and the receiver are identical and synchronized in time, where the duration of transmission of the information symbols in the carrier is a total duration of a selected number of the chips used for transmitting, and where the number of the chips is selected for the information symbols to provide a total chip energy greater than or equal to the threshold symbol energy value.

BACKGROUND OF THE INVENTION

1. Statement of the Technical Field

The invention is directed to the field of communications. In particular, the invention is directed to systems and methods for reducing bit error rates in digital chaotic spread spectrum communications systems.

2. Description of the Related Art

There are many types of communications systems known in the art, such as multiple access communications systems, low probability of intercept/low probability of detection (LPI/LPD) communications systems and spread spectrum communications systems. Many of these systems depend on constant energy symbol and/or spreading sequences. Other systems induce exploitable correlations via constant energy pulse shaping. Inherent to these constant energy pulse based waveforms, the signal energy transmitted is stationary for all practical purposes, meaning that the energy transmitted as a function of time is constant. Non-constant energy pulse spreading sequences, such as those used in coherent chaotic waveforms, have also been employed but require significantly more computational power to synchronize. However, communication signals employing non-constant energy pulse spreading sequences are typically more secure and robust to interferers.

Regardless of the type of communications system being used, one common issue in communications systems is how to minimize the bit error rate (BER). That is, what is the best method to combat errors in the recognition of individual bits in a communications stream. Bit errors are normally encountered because of the non-ideal signal transmission medium that distorts and degrades the transmitted signal; the rate of these errors decreases nonlinearly based on how the transmission channel distorts the signal and the ratio of the signal power to the noise power obtained at the receiver. In the case of digital systems, a low BER is obtained by a variety of mechanisms, including equalization, coherent combining of diverse received signals, forward error correction, interleaving, variable transmission power, and more generally transmission power margins. These measures are typically optimized when the transmitted energy per symbol is held constant. Further, there is a desire to have the capability of dynamically adjusting the symbol energy in response to a time varying environment.

SUMMARY OF THE INVENTION

Embodiments of the invention provide systems and methods for chaotic spread spectrum communications with reduced bit error rates. In a first embodiment of the invention, a method is provided for communicating a sequence of information symbols between a transmitter and a receiver using a chaotic sequence spread spectrum signal. The method includes generating identical sequences of chaotic chips at the transmitter and the receiver, the sequences of chips synchronized in time. The method also includes transmitting a sequence of information symbols in a signal using the sequence of chips generated at the transmitter, each of the information symbols in the signal having a variable duration of transmission based on a threshold symbol energy value and the chips generated at the transmitter. The method further includes extracting the sequence of information symbols from the transmitted signal, each of the information symbols in the transmitted signal extracted using the chips generated at the receiver and the threshold symbol energy value, and where the duration of transmission of each of the information symbols in the signal is a total duration of a selected number of the chips used for transmitting each information symbols, and where the number of the chips is selected for each of the information symbols to provide a total chip energy that is greater than or equal to the threshold symbol energy value.

In a second embodiment of the invention, a system for communicating a chaotic sequence spread spectrum signal is provided. The system includes a transmitter for transmitting a sequence of information symbols in a spread spectrum signal using a chaotic sequence of chips generated at the transmitter, each of the information symbols in the signal having a duration of transmission based on a threshold symbol energy value and the chips generated at the transmitter. The system also includes a receiver for extracting the sequence of information symbols from the transmitted signal, each of the information symbols in the transmitted signal extracted using a chaotic sequence of chips generated at the receiver and the threshold symbol energy value. In the system, the chaotic sequences of chips generated at the transmitter and the receiver are identical and synchronized in time, where the duration of transmission of each of the information symbols in the carrier is a total duration of a selected number of the chips used for transmitting each information symbols, and where the number of the chips is selected for each of the information symbols to provide a total chip energy that is greater than or equal to the threshold symbol energy value.

DESCRIPTION OF THE DRAWINGS

Embodiments will be described with reference to the following drawing figures, in which like numerals represent like items throughout the figures, and in which:

FIG. 1 is a conceptual diagram of a chaotic sequence generator in accordance with an embodiment of the invention.

FIG. 2 is a flow diagram of an exemplary method for generating a chaotic sequence in accordance with an embodiment of the invention.

FIG. 3 is a block diagram of an exemplary chaotic sequence generator in accordance with an embodiment of the invention.

FIG. 4 is a block diagram of a coherent chaotic spread-spectrum communications system according to an embodiment of the invention.

FIG. 5 is a block diagram of the transmitter shown in FIG. 4 according to an embodiment of the invention.

FIG. 6A is a block diagram of an embodiment of the receiver shown in FIG. 4 according to an embodiment of the invention.

FIG. 6B is a detailed block diagram of a portion of the receiver shown in FIG. 6A.

FIG. 7 is a block diagram of an embodiment of the computation device shown in FIG. 5 according to an embodiment of the invention.

DETAILED DESCRIPTION

Embodiments of the invention provide a spread-spectrum communications system employing a spreading signal generated using chaotic sequences and having improved bit error rate (BER) performance. A chaotic sequence, as that term is used herein, is a signal sequence having a time varying value expressed in a digital form that has no discernible regularity or order. In particular, the various embodiments of the invention provide a chaotic communications system in which the amount of energy per transmitted symbol is dynamically adjusted to account for the varying amplitudes of the chaotic spreading signal used to encode the data signal. That is, a communications system that adjusts the number of chips of the chaotic spreading signal used to ensure that each symbol is transmitted using at least a minimum amount of symbol energy required for reliable symbol detection at a receiver. In the various embodiments of the invention, this is accomplished by varying the length of the chaotic spreading signal used to encode a particular symbol. In other words, rather than providing a fixed symbol period length for all symbols in the data signal, a varying symbol period length is provided for each symbol being transmitted. The resulting operation that significantly constrains the energy per symbol must be coherently repeated at the receiver for accurate reception.

The invention will now be described more fully hereinafter with reference to accompanying drawings, in which illustrative embodiments of the invention are shown. This invention, may however, be embodied in many different forms and should not be construed as limited to the embodiments set forth herein. For example, the invention can be embodied as a method, a system, or a computer program product. Accordingly, the invention can take the form as an entirely hardware embodiment, an entirely software embodiment or a hardware/software embodiment.

Generation of Chaotic Sequences

One aspect of the invention provides for a digitally generated chaotic sequence for spectrally spreading data symbols by generating a sequence of chaotic chips. In this regard, it should be appreciated that the presence of any discernible pattern in a chaotic spreading sequence is much more difficult to identify as compared to patterns that emerge over time with conventional pseudo-random number sequences. As such, a chaotic spreading sequence is characterized by a greater degree of apparent randomness as compared to these conventional pseudo-random number sequences, providing a higher degree of security.

Referring now to FIG. 1, there is provided a conceptual diagram of a chaotic sequence generator 100 in accordance with the various embodiments of the invention. As shown in FIG. 1, generation of the chaotic sequence begins at processing devices 102 ₀, . . . , 102 _(N-1) where N polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) are selected. The N polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) can be selected as the same polynomial equation or as different polynomial equations. In the various embodiments of the invention, the N polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) are selected as irreducible polynomial equations having chaotic properties in Galois field arithmetic. Such irreducible polynomial equations include, but are not limited to, irreducible cubic polynomial equations and irreducible quadratic polynomial equations. The phrase “irreducible polynomial equation” as used herein refers to a polynomial equation that cannot be expressed as a product of at least two nontrivial polynomial equations over the same Galois field. For example, the polynomial equation f(x(nT)) is irreducible if there does not exist two (2) non-constant polynomial equations g(x(nT)) and h(x(nT)) in x(nT) with rational coefficients such that f(x(nT))=g(x(nT))·h(x(nT)).

As will be understood by one of ordinary skill in the art, each of the N polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) can be solved independently to obtain a respective solution. Each solution can be expressed as a residue number system (RNS) residue value using RNS arithmetic operations, i.e. modulo operations. Modulo operations are well known to one of ordinary skill in the art. Thus, such operations will not be described in great detail herein. However, it should be appreciated that a RNS residue representation for some weighted value “a” can be defined by mathematical Equation (1).

R={a modulo m₀, a modulo m₁, . . . , a modulo m_(N-1)}  (1)

where R is a RNS residue N-tuple value representing a weighted value “a”. Further, R(nT) can be a representation of the RNS solution of a polynomial equation f(x(nT)) defined as R(nT)={f₀(x(nT)) modulo m₀, f₁(x(nT)) modulo m₁, . . . , f_(N-1)(x(nT)) modulo m_(N-1)}. m₀, m₁, . . . , m_(N-1) respectively are the moduli for RNS arithmetic operations applicable to each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)).

From the foregoing, it will be appreciated that the RNS employed for solving each of the polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) respectively has a selected modulus value m₀, m₁, . . . , m_(N-1). The modulus value chosen for each RNS moduli is preferably selected to be relatively prime numbers p₀, p₁, . . . , p_(N-1). The phrase “relatively prime numbers” as used herein refers to a collection of natural numbers having no common divisors except one (1). Consequently, each RNS arithmetic operation employed for expressing a solution as an RNS residue value uses a different prime number p₀, p₁, . . . , p_(N-1) as a moduli m₀, m₁, . . . , m_(N-1).

Those of ordinary skill in the art will appreciate that the RNS residue value calculated as a solution to each one of the polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) will vary depending on the choice of prime numbers p₀, p₁, . . . , p_(N-1) selected as a moduli m₀, m₁, . . . , m_(N-1). Moreover, the range of values will depend on the choice of relatively prime numbers p₀, p₁, . . . , p_(N-1) selected as a moduli m₀, m₁, . . . , m_(N-1). For example, if the prime number five hundred three (503) is selected as modulus m₀, then an RNS solution for a first polynomial equation f₀(x(nT)) will have an integer value between zero (0) and five hundred two (502). Similarly, if the prime number four hundred ninety-one (491) is selected as modulus m₁, then the RNS solution for a second polynomial equation f₁(x(nT)) has an integer value between zero (0) and four hundred ninety (490).

According to an embodiment of the invention, each of the N polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) is selected as an irreducible cubic polynomial equation having chaotic properties in Galois field arithmetic. Each of the N polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) can also be selected to be a constant or varying function of time. The irreducible cubic polynomial equation is defined by a mathematical Equation (2).

f(x(nT))=Q(k)x ³(nT)+R(k)x ²(nT)+S(k)x(nT)+C(k,L)  (2)

where n is a sample time index value. k is a polynomial time index value. L is a constant component time index value. T is a fixed constant having a value representing a time increment. Q, R, and S are coefficients that define the polynomial equation f(x(nT)). C is a coefficient of x(nT) raised to a zero power and is therefore a constant for each polynomial characteristic. In one embodiment, a value of C is selected which empirically is determined to produce an irreducible form of the stated polynomial equation f(x(nT)) for a particular prime modulus. For a given polynomial with fixed values for Q, R, and S more than one value of C can exist, each providing a unique iterative sequence. Still, the invention is not limited in this regard.

According to another embodiment of the invention, the N polynomial equations f₀(x(nT)) . . . f_(N-1)(x(nT)) are identical exclusive of a constant value C. For example, a first polynomial equation f₀(x(nT)) is selected as f₀(x(nT))=3x³(nT)+3x²(nT)+x(nT)+C₀. A second polynomial equation f₁(x(nT)) is selected as f₁(x(nT))=3x³(nT)+3×(nT)+x(nT)+C₁. A third polynomial equation f₂(x(nT)) is selected as f₂(x(nT))=3x³(nT)+3x²(nT)+x(nT)+C₂, and so on. Each of the constant values C₀, C₁, . . . , C_(N-1) is selected to produce an irreducible form in a residue ring of the stated polynomial equation f(x(nT))=3x³(nT)+3x²(nT)+x(nT)+C. In this regard, it should be appreciated that each of the constant values C₀, C₁, . . . , C_(N-1) is associated with a particular modulus m₀, m₁, . . . , m_(N-1) value to be used for RNS arithmetic operations when solving the polynomial equation f(x(nT)). Such constant values C₀, C₁, . . . , C_(N-1) and associated modulus m₀, m₁, . . . , m_(N-1) values which produce an irreducible form of the stated polynomial equation f(x(nT)) are listed in the following Table (1).

TABLE 1 Moduli values Sets of constant values m₀, m₁, . . . , m_(N−1): C₀, C₁, . . . , C_(N−1): 3 {1, 2} 5 {1, 3} 11 {4, 9} 29 {16, 19} 47 {26, 31} 59 {18, 34} 71 {10, 19, 20, 29} 83 {22, 26, 75, 79} 101 {27, 38, 85, 96} 131 {26, 39, 77, 90} 137 {50, 117} 149 {17, 115, 136, 145} 167 {16, 32, 116, 132} 173 {72, 139} 197 {13, 96, 127, 179} 233 {52, 77} 251 {39, 100, 147, 243} 257 {110, 118} 269 {69, 80} 281 {95, 248} 293 {37, 223} 311 {107, 169} 317 {15, 55} 347 {89, 219} 443 {135, 247, 294, 406} 461 {240, 323} 467 {15, 244, 301, 425} 479 {233, 352} 491 {202, 234} 503 {8, 271} Still, the invention is not limited in this regard.

The number of discrete magnitude states (dynamic range) that can be generated with the system shown in FIG. 1 will depend on the quantity of polynomial equations N and the modulus values m₀, m₁, . . . , m_(N-1) values selected for the RNS number systems. In particular, this value can be calculated as the product M=m₀·m₁,·m₃·m₄· ·m_(N-1).

Referring again to FIG. 1, it should be appreciated that each of the RNS solutions Nos. 1 through N is expressed in a binary number system representation. As such, each of the RNS solutions Nos. 1 through N is a binary sequence of bits. Each bit of the sequence has a zero (0) value or a one (1) value. Each binary sequence has a bit length selected in accordance with a particular modulus.

According to an embodiment of the invention, each binary sequence representing a residue value has a bit length (BL) defined by a mathematical Equation (3).

BL=Ceiling[Log 2(m)]  (3)

where m is selected as one of moduli m₀, m₁, . . . , m_(N-1). Ceiling[u] refers to a next highest integer with respect to an argument u.

In order to better understand the foregoing concepts, an example is useful. In this example, six (6) relatively prime moduli are used to solve six (6) irreducible polynomial equations f₀(x(nT)), . . . , f₅(x (nT)). A prime number p₀ associated with a first modulus m₀ is selected as five hundred three (503). A prime number p₁ associated with a second modulus m₁ is selected as four hundred ninety one (491). A prime number p₂ associated with a third modulus m₂ is selected as four hundred seventy-nine (479). A prime number p₃ associated with a fourth modulus m₃ is selected as four hundred sixty-seven (467). A prime number p₄ associated with a fifth modulus m₄ is selected as two hundred fifty-seven (257). A prime number p₅ associated with a sixth modulus m₅ is selected as two hundred fifty-one (251). Possible solutions for f₀(x(nT)) are in the range of zero (0) and five hundred two (502) which can be represented in nine (9) binary digits. Possible solutions for f₁(x(nT)) are in the range of zero (0) and four hundred ninety (490) which can be represented in nine (9) binary digits. Possible solutions for f₂(x(nT)) are in the range of zero (0) and four hundred seventy eight (478) which can be represented in nine (9) binary digits. Possible solutions for f₃(x(nT)) are in the range of zero (0) and four hundred sixty six (466) which can be represented in nine (9) binary digits. Possible solutions for f₄(x(nT)) are in the range of zero (0) and two hundred fifty six (256) which can be represented in nine (9) binary digits. Possible solutions for f₅(x(nT)) are in the range of zero (0) and two hundred fifty (250) which can be represented in eight (8) binary digits. Arithmetic for calculating the recursive solutions for polynomial equations f₀(x(nT)), . . . , f₄(x (nT)) requires nine (9) bit modulo arithmetic operations. The arithmetic for calculating the recursive solutions for polynomial equation f₅(x(nT)) requires eight (8) bit modulo arithmetic operations. In aggregate, the recursive results f₀(x(nT)), . . . , f₅(x (nT)) represent values in the range from zero (0) to M−1. The value of M is calculated as follows: p₀·p₁·p₂·p₃·p₄·p₅=503·491·479·467·257·251=3,563,762,191,059,523. The binary number system representation of each RNS solution can be computed using Ceiling[Log 2(3,563,762,191,059,523)]=Ceiling[51.66]=52 bits. Because each polynomial is irreducible, all 3,563,762,191,059,523 possible values are computed resulting in a sequence repetition time of M times T seconds, i.e, a sequence repetition times an interval of time between the computation of each values in the sequence of generated values. Still, the invention is not limited in this regard.

Referring again to FIG. 1, the generation of a chaotic sequence continues with mapping operation performed by a mapping device 104. The mapping operations involve mapping the RNS solutions Nos. 1 through N to a weighted number system representation to form a chaotic sequence output. The phrase “weighted number system” as used herein refers to a number system other than a residue number system. Such weighted number systems include, but are not limited to, an integer number system, a binary number system, an octal number system, and a hexadecimal number system.

In some embodiments of the invention, the RNS solutions Nos. 1 through N are mapped to a weighted number system representation by determining a series of digits in the weighted number system based on the RNS solutions Nos. 1 through N. The term “digit” as used herein refers to a symbol of a combination of symbols to represent a number. For example, a digit can be a particular bit of a binary sequence. In other embodiments of the invention, the RNS solutions Nos. 1 through N are mapped to a weighted number system representation by identifying a number in the weighted number system that is defined by the RNS solutions Nos. 1 through N. According to yet another embodiment of the invention, the RNS solutions Nos. 1 through N are mapped to a weighted number system representation by identifying a truncated portion of a number in the weighted number system that is defined by the RNS solutions Nos. 1 through N. The truncated portion can include any serially arranged set of digits of the number in the weighted number system. The truncated portion can also be exclusive of a most significant digit of the number in the weighted number system. The phrase “truncated portion” as used herein refers to a chaotic sequence with one or more digits removed from its beginning and/or ending. The phrase “truncated portion” also refers to a segment including a defined number of digits extracted from a chaotic sequence. The phrase “truncated portion” also refers to a result of a partial mapping of the RNS solutions Nos. 1 through N to a weighted number system representation.

In some embodiments of the invention, a mixed-radix conversion method is used for mapping RNS solutions Nos. 1 through N to a weighted number system representation. “The mixed-radix conversion procedure to be described here can be implemented in” [modulo moduli only and not modulo the product of moduli.] See Residue Arithmetic and Its Applications To Computer Technology, written by Nicholas S. Szabo & Richard I. Tanaka, McGraw-Hill Book Co., New York, 1967. [In a mixed-radix number system,] “a number x may be expressed in a mixed-radix form:

$\begin{matrix} {x = {{a_{N}{\prod\limits_{i = 1}^{N - 1}\; R_{i}}} + \ldots + {a_{3}R_{1}R_{2}} + {a_{2}R_{1}} + a_{1}}} & (4) \end{matrix}$

where the R_(i) are the radices, the a_(i) are the mixed-radix digits, and 0≦a_(i)<R_(i). For a given set of radices, the mixed-radix representation of x is denoted by (a_(n), a_(n-1), . . . , a₁) where the digits are listed order of decreasing significance.” See Id. “The multipliers of the digits a_(i) are the mixed-radix weights where the weight of a_(i) is

$\begin{matrix} {{{{\prod\limits_{j = 1}^{i - 1}\; {R_{j}\mspace{14mu} {for}\mspace{14mu} i}} \neq 1.}"}\mspace{14mu} {See}\mspace{14mu} {{Id}.}} & (5) \end{matrix}$

For conversion from the RNS to a mixed-radix system, a set of moduli are chosen so that m_(i)=R_(i). A set of moduli are also chosen so that a mixed-radix system and a RNS are said to be associated. “In this case, the associated systems have the same range of values, that is

$\begin{matrix} {\prod\limits_{i = 1}^{N}\; {m_{i}.}} & (6) \end{matrix}$

The mixed-radix conversion process described here may then be used to convert from the [RNS] to the mixed-radix system.” See Id.

“If m_(i)=R_(i), then the mixed-radix expression is of the form:

$\begin{matrix} {x = {{a_{N}{\prod\limits_{i = 1}^{N - 1}\; m_{i}}} + \ldots + {a_{3}m_{1}m_{2}} + {a_{2}m_{1}} + a_{1}}} & (7) \end{matrix}$

where a_(i) are the mixed-radix coefficients. The a_(i) are determined sequentially in the following manner, starting with a₁.” See Id.

$\begin{matrix} {x = {{a_{N}{\prod\limits_{i = 1}^{N - 1}\; m_{i}}} + \ldots + {a_{3}m_{1}m_{2}} + {a_{2}m_{1}} + a_{1}}} & (8) \end{matrix}$

is first taken modulo m₁. “Since all terms except the last are multiples of m₁, we have

x

_(m) ₁ =a₁. Hence, a₁ is just the first residue digit.” See Id.

“To obtain a₂, one first forms x-a₁ in its residue code. The quantity x-a₁ is obviously divisible by m₁. Furthermore, m₁ is relatively prime to all other moduli, by definition. Hence, the division remainder zero procedure [Division where the dividend is known to be an integer multiple of the divisor and the divisor is known to be relatively prime to M] can be used to find the residue digits of order 2 through N of

$\frac{x - a_{1}}{m_{1}}.$

Inspection of

$\begin{matrix} \left\lbrack {x = {{a_{N}{\prod\limits_{i = 1}^{N - 1}\; m_{i}}} + \ldots + {a_{3}m_{1}m_{2}} + {a_{2}m_{1}} + a_{1}}} \right\rbrack & (9) \end{matrix}$

shows then that x is a₂. In this way, by successive subtracting and dividing in residue notation, all of the mixed-radix digits may be obtained.” See Id.

“It is interesting to note that

$\begin{matrix} {{a_{1} = {\langle x\rangle}_{m_{1}}},{a_{2} = {\langle\left\lfloor \frac{x}{m_{1}} \right\rfloor\rangle}_{m_{2}}},{a_{3} = {\langle\left\lfloor \frac{x}{m_{1}m_{2}} \right\rfloor\rangle}_{m_{3}}}} & (10) \end{matrix}$

and in general for i>1

$\begin{matrix} {a_{i} = {\langle\left\lfloor \frac{x}{m_{1}m_{2}\mspace{14mu} \ldots \mspace{14mu} m_{i - 1}} \right\rfloor\rangle}_{m_{i}}} & (11) \end{matrix}$

.” See Id. From the preceding description it is seen that the mixed-radix conversion process is iterative. The conversion can be modified to yield a truncated result. Still, the invention is not limited in this regard.

In some embodiments of the invention, a Chinese remainder theorem (CRT) arithmetic operation is used to map the RNS solutions Nos. 1 through N to a weighted number system representation. The CRT arithmetic operation can be defined by a mathematical Equation (12).

$\begin{matrix} {{Y({nT})} = {\langle\begin{matrix} {{{\langle{\left\lbrack {{3{x_{0}^{3}({nT})}} + {3{x_{0}^{2}({nT})}} + {x_{0}({nT})} + C_{0}} \right\rbrack b_{0}}\rangle}_{p_{0}}\frac{M}{p_{0}}} + \ldots +} \\ {{\langle{\left\lbrack {{3{x_{N - 1}^{3}({nT})}} + {3{x_{N - 1}^{2}({nT})}} + {x_{N - 1}({nT})} + C_{N - 1}} \right\rbrack b_{N - 1}}\rangle}_{p_{N - 1}}\frac{M}{p_{N - 1}}} \end{matrix}\rangle}_{M}} & (12) \end{matrix}$

Equation (12) can be re-written in iterative form as mathematical Equation (12a).

$\begin{matrix} {{Y({nT})} = {\langle\begin{matrix} {{{\langle{\begin{bmatrix} {{3{x_{0}^{3}\left( {\left( {n - 1} \right)T} \right)}} + {3x_{0}^{2}\left( {\left( {n - 1} \right)T} \right)} +} \\ {{x_{0}\left( {\left( {n - 1} \right)T} \right)} + C_{0}} \end{bmatrix}b_{0}}\rangle}_{p_{0}}\frac{M}{p_{0}}} + \ldots +} \\ {{\langle{\begin{bmatrix} {{3{x_{N - 1}^{3}\left( {\left( {n - 1} \right)T} \right)}} + {3x_{N - 1}^{2}\left( {\left( {n - 1} \right)T} \right)} +} \\ {{x_{N - 1}\left( {\left( {n - 1} \right)T} \right)} + C_{N - 1}} \end{bmatrix}b_{N - 1}}\rangle}_{p_{N - 1}}\frac{M}{p_{N - 1}}} \end{matrix}\rangle}_{M}} & \left( {12a} \right) \end{matrix}$

where Y(nT) is the result of the CRT arithmetic operation. n is a sample time index value. T is a fixed constant having a value representing a time interval or increment. x₀-x_(N-1) are RNS solutions Nos. 1 through N. p₀, p₁, . . . , p_(N-1) are prime number moduli. M is a fixed constant defined by a product of the relatively prime numbers p₀, p₁, p_(N-1). b₀, b₁, . . . , b_(N-1) are fixed constants that are chosen as the multiplicative inverses of the product of all other primes modulo p₀, p₁, . . . , p_(N-1), respectively. Equivalently,

$\begin{matrix} {b_{j} = {\left( \frac{M}{p_{j}} \right)^{- 1}{mod}\; {p_{j}.}}} & (13) \end{matrix}$

The b_(j)'s enable an isomorphic and equal mapping between an RNS N-tuple value representing a weighted number and said weighted number. However without loss of chaotic properties, the mapping need only be unique and isomorphic. As such, a weighted number x can map into a tuple y. The tuple y can map into a weighted number z. The weighted number x is not equal to x as long as all tuples map into unique values for z in a range from zero (0) to M−1. Therefore, in some embodiments of the invention, the b_(j)'s can be defined as

$\begin{matrix} {b_{j} = {\left( \frac{M}{p_{j}} \right)^{- 1}{mod}\; {p_{j}.}}} & (14) \end{matrix}$

In other embodiments of the invention, all b_(j)'s can be set equal to one or more values without loss of the chaotic properties. Different values of b_(j) apply a bijective mapping within the RNS, but do not interfere with the CRT combination process.

The chaotic sequence output Y(nT) can be expressed in a binary number system representation. As such, the chaotic sequence output Y(nT) can be represented as a binary sequence. Each bit of the binary sequence has a zero (0) value or a one (1) value. The chaotic sequence output Y(nT) can have a maximum bit length (MBL) defined by a mathematical Equation (15).

MBL=Ceiling[Log 2(M)]  (15)

where M is the product of the relatively prime numbers p₀, p₁, p_(N-1) selected as moduli m₀, m₁, . . . , m_(N-1). In this regard, it should be appreciated the M represents a dynamic range of a CRT arithmetic operation. The phrase “dynamic range” as used herein refers to a maximum possible range of outcome values of a CRT arithmetic operation. Accordingly, the CRT arithmetic operation generates a chaotic numerical sequence with a periodicity equal to the inverse of the dynamic range M. The dynamic range requires a Ceiling[Log 2(M)] bit precision.

In some embodiments of the invention, M equals three quadrillion five hundred sixty-three trillion seven hundred sixty-two billion one hundred ninety-one million fifty-nine thousand five hundred twenty-three (3,563,762,191,059,523). By substituting the value of M into Equation (6), the bit length (BL) for a chaotic sequence output Y expressed in a binary system representation can be calculated as follows: BL=Ceiling/Log 2(3,563,762,191,059,523)=52 bits. As such, the chaotic sequence output Y(nT) is a fifty-two (52) bit binary sequence having an integer value between zero (0) and three quadrillion five hundred sixty-three trillion seven hundred sixty-two billion one hundred ninety-one million fifty-nine thousand five hundred twenty-two (3,563,762,191,059,522), inclusive. Still, the invention is not limited in this regard. For example, the chaotic sequence output Y(nT) can be a binary sequence representing a truncated portion of a value between zero (0) and M−1. In such a scenario, the chaotic sequence output Y(nT) can have a bit length less than Ceiling[Log 2(M)]. It should be noted that while truncation affects the dynamic range of the system it has no effect on the periodicity of a generated sequence.

As one of ordinary skill in art will recognize, the above-described chaotic sequence generation can be iteratively performed. In such a scenario, a feedback mechanism (e.g., a feedback loop) can be provided so that a variable “x” of a polynomial equation can be selectively defined as a solution computed in a previous iteration. Mathematical Equation (2) can be rewritten in a general iterative form: f(x(nT)=Q(k)x³((n−1)T)+R(k)x²((n−1)T)+S(k)x((n−1)T)+C(k,L). For example, a fixed coefficient polynomial equation is selected as f(x(n·1 ms))=3x³((n−1)·1 ms)+3x²((n−1)·1 ms)+x((n−1)·1 ms)+8 modulo 503. n is a variable having a value defined by an iteration being performed. x is a variable having a value allowable in a residue ring. In a first iteration, n equals one (1) and x is selected as two (2) which is allowable in a residue ring. By substituting the value of n and x into the stated polynomial equation f(x(nT)), a first solution having a value forty-six one (46) is obtained. In a second iteration, n is incremented by one and x equals the value of the first solution, i.e., forty-six (46) resulting in the solution 298, 410 mod 503 or one hundred thirty-one (131). In a third iteration, n is again incremented by one and x equals the value of the second solution.

Referring now to FIG. 2, there is provided a flow diagram of an exemplary method 200 for generating a chaotic sequence according to an embodiment of the invention. As shown in FIG. 2, the method 200 begins with step 202 and continues with step 204. In step 204, a plurality of polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) are selected. In this regard, it should be appreciated that the polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) can be selected as the same polynomial equation except for a different constant term or different polynomial equations. After step 204, step 206 is performed where a determination for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) is made as to which combinations of RNS moduli m₀, m₁, . . . , m_(N-1) used for arithmetic operations and respective constant values C₀, C₁, . . . , C_(N-1) generate irreducible forms of each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)). In step 208, a modulus is selected for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) that is to be used for RNS arithmetic operations when solving the polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)). In this regard, it should be appreciated that the modulus is selected from the moduli identified in step 206. It should also be appreciated that a different modulus must be selected for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)).

As shown in FIG. 2, the method 200 continues with step 210. In step 210, a constant C_(m) is selected for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) for which a modulus is selected. Each constant C_(m) corresponds to the modulus selected for the respective polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)). Each constant C_(m) is selected from among the possible constant values identified in step 206 for generating an irreducible form of the respective polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)).

After step 210, the method 200 continues with step 212. In step 212, a value for time increment “T” is selected. Thereafter, an initial value for “x” is selected. In this regard, it should be appreciated that the initial value for “x” can be any value allowable in a residue ring. Subsequently, step 216 is performed where RNS arithmetic operations are used to iteratively determine RNS solutions for each of the stated polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)). In step 218, a series of digits in a weighted number system are determined based in the RNS solutions. This step can involve performing a mixed radix arithmetic operation or a CRT arithmetic operation using the RNS solutions to obtain a chaotic sequence output.

After step 218, the method 200 continues with a decision step 220. If a chaos generator is not terminated (220:NO), then step 224 is performed where a value of “x” in each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) is set equal to the RNS solution computed for the respective polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) in step 216. Subsequently, the method 200 returns to step 216. If the chaos generator is terminated (220:YES), then step 222 is performed where the method 200 ends.

One of ordinary skill in the art will appreciate that the method 200 is only one exemplary method for generating a chaotic sequence. However, the invention is not limited in this regard and any other method for generating a chaotic sequence can be used without limitation.

Referring now to FIG. 3, there is illustrated an exemplary chaotic sequence generator 300 in accordance with an embodiment of the invention. The chaotic sequence generator 300 is comprised of hardware and/or software configured to generate a digital chaotic sequence. In this regard, it should be appreciated that the chaotic sequence generator 300 is comprised of computing processors 302 ₀-302 _(N-1). The chaotic sequence generator 300 is also comprised of a mapping processor 304. Each computing processor 302 ₀-302 _(N-1) is coupled to the mapping processor 304 by a respective data bus 306 ₀-306 _(N-1). As such, each computing processor 302 ₀-302 _(N-1) is configured to communicate data to the mapping processor 304 via a respective data bus 306 ₀-306 _(N-1). The mapping processor 304 can be coupled to an external device (not shown) via a data bus 308. In this regard, it should be appreciated that the external device (not shown) includes, but is not limited to, a communication device configured to combine or modify a signal in accordance with a chaotic sequence output.

Referring again to FIG. 3, the computing processors 302 ₀-302 _(N-1) are comprised of hardware and/or software configured to solve N polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) to obtain a plurality of solutions. The N polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) can be irreducible polynomial equations having chaotic properties in Galois field arithmetic. Such irreducible polynomial equations include, but are not limited to, irreducible cubic polynomial equations and irreducible quadratic polynomial equations. The N polynomial equations f₀(x(nT)) . . . f_(N-1)(x(nT)) can also be identical exclusive of a constant value. The constant value can be selected so that a polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) is irreducible for a predefined modulus. The N polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) can further be selected as a constant or varying function of time.

Each of the solutions can be expressed as a unique residue number system (RNS) N-tuple representation. In this regard, it should be appreciated that the computing processors 302 ₀-302 _(N-1) employ modulo operations to calculate a respective solution for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) using modulo based arithmetic operations. Each of the computing processors 302 ₀-302 _(N-1) are comprised of hardware and/or software configured to utilize a different relatively prime number p₀, p₁, . . . , p_(N-1) as a moduli m₀, m₁, . . . , m_(N-1) for modulo based arithmetic operations. The computing processors 302 ₀-302 _(N-1) are also comprised of hardware and/or software configured to utilize modulus m₀, m₁, . . . , m_(N-1) selected for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) so that each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) is irreducible. The computing processors 302 ₀-302 _(N-1) are further comprised of hardware and/or software configured to utilize moduli m₀, m₁, . . . , m_(N-1) selected for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) so that solutions iteratively computed via a feedback mechanism 310 ₀-310 _(N-1) are chaotic. In this regard, it should be appreciated that the feedback mechanisms 310 ₀-310 _(N-1) are provided so that the solutions for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) can be iteratively computed. Accordingly, the feedback mechanisms 310 ₀-310 _(N-1) are comprised of hardware and/or software configured to selectively define a variable “x” of a polynomial equation as a solution computed in a previous iteration.

Referring again to FIG. 3, the computing processors 302 ₀-302 _(N-1) are further comprised of hardware and/or software configured to express each of the RNS residue values in a binary number system representation. In this regard, the computing processors 302 ₀-302 _(N-1) can employ an RNS-to-binary conversion method. Such methods are generally known to one of ordinary skill in the art and therefore will not be described in great detail herein. However, it should be appreciated that any such method can be used without limitation. It should also be appreciated that the residue values expressed in binary number system representations are hereinafter referred to as moduli solutions Nos. 1 through N comprising the elements of an RNS N-tuple.

According to an embodiment of the invention, the computing processors 302 ₀-302 _(N-1) are further comprised of memory based tables (not shown) containing pre-computed residue values in a binary number system representation. The address space of each memory table is at least from zero (0) to m_(m)−1 for all m, m₀ through m_(N-1). On each iteration, the table address is used to initiate the sequence. Still, the invention is not limited in this regard.

Referring again to FIG. 3, the mapping processor 304 is comprised of hardware and/or software configured to map the moduli (RNS N-tuple) solutions Nos. 1 through N to a weighted number system representation. The result is a series of digits in the weighted number system based on the moduli solutions Nos. 1 through N. For example, the mapping processor 304 can be comprised of hardware and/or software configured to determine the series of digits in the weighted number system based on the RNS residue values using a Chinese Remainder Theorem process. In this regard, it will be appreciated by one of ordinary skill in the art that the mapping processor 304 is comprised of hardware and/or software configured to identify a number in the weighted number system that is defined by the moduli solutions Nos. 1 through N.

In the various embodiments of the invention, the mapping processor 304 can be comprised of hardware and/or software configured to identify a truncated portion of a number in the weighted number system that is defined by the moduli solutions Nos. 1 through N. For example, the mapping processor 304 can also be comprised of hardware and/or software configured to select the truncated portion to include any serially arranged set of digits of the number in the weighted number system. Further, the mapping processor 304 can include hardware and/or software configured to select the truncated portion to be exclusive of a most significant digit when all possible weighted numbers represented by P bits are not mapped, i.e., when M−1<2^(P). P is a fewest number of bits required to achieve a binary representation of the weighted numbers. Still, the invention is not limited in this regard.

Referring again to FIG. 3, the mapping processor 304 is comprised of hardware and/or software configured to express a chaotic sequence in a binary number system representation. In this regard, it should be appreciated that the mapping processor 304 can employ a weighted-to-binary conversion method. Such methods are generally known to one of ordinary skill in the art and therefore will not be described in great detail herein. However, it should be appreciated that any such method can be used without limitation.

One of ordinary skill in the art will appreciate that the chaotic generator 300 shown in FIG. 3 is an exemplary architecture for a chaotic generator. However, the invention is not limited in this regard and any other chaotic generator architecture can be used without limitation.

Spread Spectrum Communications with Chaotic Sequences

As described above, another aspect of the invention provides for using a communication system disclosed that utilizes a spread spectrum communications system using chaotic sequences, hereinafter a coherent chaotic sequence spread spectrum (CCSSS) method. That is, prior to being transmitted, symbols in a data signal are combined with a higher rate chaotic sequence (analogous to the binary PN spreading sequence known as a chipping code in conventional direct sequence spread spectrum systems) that spreads the spectrum of the data according to a spreading ratio. The resulting signal resembles a truly random signal, but this randomness can be deciphered at the receiving end to recover the original data. In particular, the data signal is recovered by despreading the received signal using the same chaotic sequence which is generated coherently at a receiver. The CCSSS system in relation to FIGS. 4 through 6B channel encodes a baseband carrier with PSK symbols. The CCSSS system also modulates the phase modulated carrier in a chaotic manner utilizing a string of discrete time chaotic samples. The discrete time chaotic samples shall hereinafter be referred to as “chips”. As will be appreciated by those familiar with direct sequence spread spectrum (DSSS) systems, each chip will generally have a much shorter time interval than the duration of each of the information symbols in the data signal. Thus it will be understood that the carrier is modulated using the chaotic sequence chips. Moreover, it will be understood that the chip rate associated with the chaotic sequence is much higher than the symbol rate in the data signal. It should also be understood that the chaotic sequence of chips which are utilized for generating the transmitted signal is known a priori by the receiver. Consequently, the same chaotic sequence can be used at the receiver to reconstruct the data signal or remove the effect of spreading at the receiver.

Referring now to FIG. 4, there is provided a coherent chaotic spread-spectrum communication system 400 according to an embodiment of the invention. The coherent chaotic spread-spectrum communication system 400 is comprised of a transmitter 402 and a receiver 404. The transmitter 402 is configured to generate an amplitude-and-time-discrete baseband signal and to spread the amplitude-and-time-discrete baseband signal over a wide intermediate frequency band. This spreading consists of multiplying the amplitude-and-time-discrete baseband signal by a digital chaotic sequence. The product of this arithmetic operation is hereinafter referred to as a digital chaotic signal. In this regard, it should be understood that the transmitter 402 is also configured to process the digital chaotic signal to place the same in a proper analog form suitable for transmission over a communications link. The transmitter 402 is further configured to communicate or transmit analog chaotic signals 406 to the receiver 404 via a communications link. The transmitter 402 will be described in greater detail below in relation to FIG. 5.

The receiver 404 is configured to receive transmitted analog chaotic signals 406 from the transmitter 402. The receiver 404 is also configured to down convert, digitize, and de-spread the analog chaotic signals 406 by correlating it with a replica of the chaotic sequence generated at the transmitter 402. The chaotic sequence is also time synchronized to the analog chaotic signal 406: i.e., a sampling rate of the chaotic sequence used in the receiver is synchronized with a clock (not shown) of the transmitter 402. The output of the arithmetic operation that de-spreads the analog chaotic signal 406 is hereinafter referred to as a de-spread signal. In this regard, it should be understood that the receiver 404 is further configured to process a de-spread signal for obtaining data contained therein. The receiver 404 is configured to convert the data into text, sound, pictures, navigational-position information, and/or any other type of useful payload information that can be communicated. The receiver 404 is described in greater detail below in relation to FIGS. 6A and 6B.

Referring now to FIG. 5, there is provided a block diagram of the transmitter 402 shown in FIG. 4 that is useful for understanding the invention. It should be noted that the embodiment of FIG. 5 assumes that: (1) a low order phase shift keying (PSK) data modulation is used; (2) no pulse shaping is applied to information symbols; (3) channel encoded symbols are generated in quadrature form; and (4) chaotic spectral spreading is performed at an intermediate frequency (IF).

Referring again to FIG. 5, the transmitter 402 is comprised of a data source 502. The transmitter 402 is also comprised of a source encoder 504, a symbol data formatter 506, an acquisition data generator 508, a transmitter controller 510, a multiplexer 514, a channel encoder 516, a precision real time reference 512, and a digital complex multiplier 524. The transmitter 402 is further comprised of a chaos generator 300, a real uniform statistics to quadrature Gaussian statistics mapper device (RUQG) 520, and a sample rate change filter (SRCF) 522. The transmitter 402 is further comprised of an interpolator 526, a digital local oscillator (LO) 530, a real part of a complex multiplier 528, a digital-to-analog converter (DAC) 532, an anti-image filter 534, an intermediate frequency (IF) to radio frequency (RF) conversion device 536, and an antenna element 538. Each of the above listed components 502-516, 520-538 are well known to one of ordinary skill in the art. Thus, these components will not be described in great detail herein. However, a brief discussion of the transmitter 402 architecture is provided to assist a reader in understanding the invention.

Referring again to FIG. 5, the data source 502 is configured to receive bits of data from an external data source (not shown) as bits of data. In this regard, it should be appreciated that the data source 502 is an interface configured for receiving an input signal containing data from an external device (not shown). The data source 502 is further configured to supply bits of data to the source encoder 504 at a particular data transfer rate. The source encoder 504 can be configured to encode the data received from the external device (not shown) using a forward error correction coding scheme. The bits of data received at or generated by the source encoder 504 represent any type of information that may be of interest to a user. For example, the data can be used to represent text, telemetry, audio, or video data. The source encoder 504 is further configured to supply bits of data to the symbol data formatter 506 at a particular data transfer rate.

The symbol data formatter 506 is configured to process bits of data for forming channel encoded symbols. In one embodiment of the invention, the source encoded symbols are phase shift keyed (PSK) encoded. If it is desired to use a non-coherent form of PSK with the coherent chaos spread spectrum system, then the symbol data formatter 506 can also be configured to differentially encode formed PSK symbol data words. Differential encoding is well known to one of ordinary skill in the art and therefore will not be described in great detail herein. The symbol data formatter 506 can be further configured to communicate non-differentially encoded PSK symbol data words and/or differentially encoded PSK symbol data words to the multiplexer 514. Still, the invention is not limited in this regard.

In one embodiment of the invention, the symbol data formatter 506 is functionally similar to a serial in/parallel out shift register where the number of parallel bits out is equal to log base two (log₂) of the order of the channel encoder 516. In this regard, the symbol data formatter 506 is selected for use with a quadrature phase shift keying (QPSK) channel encoder. As such, the symbol data formatter 506 is configured to perform a QPSK formatting function for grouping two (2) bits of data together to form a QPSK symbol data word (i.e., a single two bit parallel word). Thereafter, the symbol data formatter 506 communicates the symbol data word to the multiplexer 514. Still, the invention is not limited in this regard.

In another embodiment of the invention, the symbol data formatter 506 is functionally similar to a serial in/parallel out shift register where the number of parallel bits out is equal to log base two (log₂) of the order of the channel encoder 516. In this regard, the symbol data formatter 506 is selected for use with a binary phase shift keying (BPSK) channel encoder. As such, the symbol data formatter 506 is configured to map one bit of data per symbol time to a BPSK channel encoder. Thereafter, the symbol data formatter 506 communicates the BPSK symbol data word to the multiplexer 514. Still, the invention is not limited in this regard.

In still another embodiment of the invention, the symbol data formatter 506 is selected for use with a sixteen quadrature amplitude modulation (16 QAM) modulator. As such, the symbol data formatter 506 is configured to map four (4) bits to a 16 QAM symbol data word. Thereafter, the symbol data formatter 506 communicates the 16 QAM symbol data word to the multiplexer 514. Still, the invention is not limited in this regard.

In yet another embodiment of the invention, the symbol data formatter 506 is selected for use with a binary amplitude shift keying (ASK) modulator. As such, the symbol data formatter 506 is configured to map one bit of data to a ASK symbol data word. Thereafter, the symbol data formatter 506 communicates the ASK symbol data word to the multiplexer 514. Still, the invention is not limited in this regard.

The transmitter 402 also includes an acquisition data generator 508 capable of generating a “known data preamble” that can be used to enable initial synchronization of a chaotic sequence generated in the transmitter 402 and the receiver 404. The duration of this “known data preamble” is determined by an amount required by the receiver 404 to synchronize with the transmitter 402 under known worst case channel conditions. In some embodiments of the invention, the “known data preamble” is a repetition of the same known symbol. In other embodiments of the invention, the “known data preamble” is a series of known symbols. The acquisition data generator 508 can be further configured to communicate the “known data preamble” to the multiplexer 514.

Referring again to FIG. 5, the multiplexer 514 is configured to receive the binary word to be modulated by the channel encoder from the symbol data formatter 506. The multiplexer 514 is also configured to receive a “known data preamble” from the acquisition data generator 508. The multiplexer 514 is coupled to the transmitter controller 510. The transmitter controller 510 is configured to control the multiplexer 514 so that the multiplexer 514 routes the “known data preamble” to the channel encoder 516 at the time of a new transmission.

In some embodiments of the invention, the “known data preamble” is stored in a modulated form. In such embodiments, the architecture of FIG. 5 is modified such that the multiplexer 514 exists after the channel encoder 516. Still, the invention is not limited in this regard.

According to another embodiment of the invention, the “known data preamble” may be injected at known intervals to aid in periodic resynchronization of the chaotic sequence generated in the transmitter 402 and the receiver 404. This would typically be the case for an implementation meant to operate in harsh channel conditions. Still, the invention is not limited in this regard.

Referring again to FIG. 5, the multiplexer 514 is configured to select one or more information symbol words to be routed to the channel encoder 516 after a preamble period has expired. The multiplexer 514 is also configured to communicate the information symbol words to the channel encoder 516. In this regard, it should be appreciated that a communication of the information symbol data words to the channel encoder 516 is delayed by a time defined by the length of the “known data preamble.” As should be appreciated, this delay allows all of a “known data preamble” to be fully communicated to the channel encoder 516 prior to communication of the information symbols.

Referring again to FIG. 5, the channel encoder 516 is configured to perform actions for representing the “known data preamble” and the information symbol data words in the form of a modulated amplitude-and-time-discrete digital signal. The modulated amplitude-and-time-discrete digital signal is defined by digital words which represent intermediate frequency (IF) channel encoded symbols comprised of bits of data having a one (1) value or a zero (0) value. Methods for representing digital symbols by an amplitude-and-time-discrete digital signal are well known to one of ordinary skill in the art. Thus, such methods will not be described in great detail herein. However, it should be appreciated that the channel encoder 516 can employ any such method. For example, the channel encoder 516 can be selected as a digital baseband modulator employing quadrature phase shift keying (QPSK). As will be appreciated by one of ordinary skill in the art, the output of the QPSK channel encoder will include an in-phase (“I”) data and quadrature phase (“Q”) data. The I and Q data will be thereafter communicated to the digital complex multiplier 524. However, the invention is not limited in this regard.

According to an embodiment of the invention, the transmitter 402 is further comprised of a sample rate matching device (not shown) between the channel encoder 516 and the digital complex multiplier 524. The sample rate matching device (not shown) is provided for holding the amplitude-and-time-discrete digital signal for a duration compatible with the chaos sampling rate. As should be appreciated, the sample rate matching device (not shown) performs sample duration matching on the amplitude-and-time-discrete digital signal so that a sample duration of the amplitude-and-time-discrete digital signal is consistent with a digital chaotic sequence communicated to the digital complex multiplier 524. Still, the invention is not limited in this regard.

Referring again to FIG. 5, the digital complex multiplier 524 performs a complex multiplication in the digital domain. In the digital complex multiplier 524, the amplitude-and-time-discrete digital signal from the channel encoder 516 is multiplied by a digital representation of a chaotic sequence. The chaotic sequence is generated in the chaos generator 300, as described with reference to FIG. 3. Thus, the description provided above in relation to FIG. 3 is sufficient for understanding the architecture of chaos generator 300 shown in FIG. 5.

The rate at which the digital chaotic sequence is generated is an integer multiple of an information symbol rate. The greater the ratio between the information symbol period and the sample period of the digital chaotic sequence, the higher a spreading gain. The chaos generator 300 communicates the chaotic sequence to a RUQG 520. The RUQG 520 is configured to statistically transform a digital chaotic sequence into a transformed digital chaotic sequence with pre-determined statistical properties. The transformed digital chaotic sequence can have a characteristic form including combinations of real, complex, or quadrature, being of different word widths, and having different statistical distributions. For example, the RUQG 520 may take in two (2) uniformly distributed real inputs from the chaos generator 300 and convert those via a complex-valued bivariate Box-Muller transformation to a quadrature output having statistical characteristics of a Gaussian distribution. Such conversions are well understood by one of ordinary skill in the art, and therefore will not be described in great detail herein. However, it should be understood that such techniques may use nonlinear processors, look-up tables, iterative processing (CORDIC functions), or other similar mathematical processes. The RUQG 520 is further configured to communicate transformed chaotic sequences to the SRCF 522.

The statistically transformed output of the digital chaotic sequence has a multi-bit resolution consistent with a resolution of the DAC 532. The RUQG 520 communicates the statistically transformed output of the digital chaotic sequence to the SRCF 522. For example, the RUQG 520 communicates an in-phase (“I”) data and quadrature phase (“Q”) data to the SRCF 522 when the channel encoder 516 is configured to yield a complex output representation. Still, the invention is not limited in this regard.

If the chaos sample rate of the transformed chaotic sequence is different than the sample rate required by subsequent signal processing, then the two rates must be matched. The chaotic sequence can therefore be resampled in the SRCF 522. For example, SRCF 522 can be comprised of a real interpolation filters to upsample each of the in-phase and quadrature-phase processing paths of the chaotic sequence. As should be appreciated, the SRCF 522 performs a sample rate change on the transformed digital chaotic sequence so that a sample rate of the transformed digital chaotic sequence is the same as the sampling rates required by subsequent signal processing operations. The SRCF 522 is also configured to communicate a resampled, transformed digital chaotic sequence to the digital complex multiplier 524.

According to an embodiment of the invention, the RUQG 520 statistically transforms a digital chaotic sequence into a quadrature Gaussian form of the digital chaotic sequence. This statistical transformation is achieved via a nonlinear processor that combines lookup tables and embedded computational logic to implement the conversion of two (2) independent uniformly distributed random variables into a quadrature pair of Gaussian distributed variables. One such structure for this conversion is as shown in the mathematical expressions (16) and (17).

G ₁=√{square root over (−2 log(u ₁))}·cos(2πu ₂)  (16)

G ₂=√{square root over (−2 log(u ₁))}·sin(2πu ₂)  (17)

where {u1, u2} are uniformly distributed independent input random variables and {G₁, G₂} are Gaussian distributed output random variables. In such a scenario, the SRCF 522 is comprised of one sample rate change filter to resample an in-phase (“I”) data sequence and a second sample rate change filter to resample a quadrature-phase (“Q”) data sequence. The SRCF 522 is configured to communicate a resampled, transformed digital chaotic sequence to the digital complex multiplier 524. More particularly, the SRCF 522 communicates an in-phase (“I”) data and quadrature phase (“Q”) data to the digital complex multiplier 524. Still, the invention is not limited in this regard.

The digital complex multiplier 524 performs a complex multiplication on the digital chaotic sequence output from the SRCF 522 and the amplitude-and-time-discrete digital signal output from the channel encoder 516. The resulting output is a digital representation of a chaotic sequence spread spectrum modulated IF signal in which the digital data from the channel encoder 516 has been spread over a wide frequency bandwidth in accordance with a chaotic sequence generated by the chaos generator 300.

The digital complex multiplier 524 is configured to combine a digital chaotic sequence with an amplitude-and-time-discrete digital signal using an arithmetic operation. The arithmetic operation is selected as a complex-valued digital multiplication operation. The complex-valued digital multiplication operation includes multiplying the amplitude-and-time-discrete digital signal by the digital chaotic sequence to obtain a digital chaotic output signal. The digital complex multiplier 524 is also configured to communicate digital chaotic output signals to the interpolator 526.

The interpolator 526, real part of complex multiplier 528 and quadrature digital local oscillator 530 operate in tandem to form an intermediate frequency (IF) translator which frequency modulates a quadrature first intermediate frequency (IF) signal received from the complex multiplier to a second real intermediate frequency (IF) signal. Such digital intermediate frequency (IF) translators are known to one of ordinary skill in the art and shall not be discussed in detail here.

The interpolator 526 accepts an input from the complex multiplier 524. In one embodiment the modulated symbols are in quadrature form and the interpolator is implemented as two real interpolators. Still, the invention is not limited in this regard.

The interpolator 526 raises the sample rate of the amplitude-and-time-discrete digital signal received from the complex multiplier 524 to a rate compatible with the bandwidth and center frequency of the second IF. The digital local oscillator 530 generates a complex quadrature amplitude-and-time-discrete digital sinusoid at a frequency which shall translate the first intermediate frequency (IF) to a desired second intermediate frequency (IF). The digital local oscillator 530 is also configured to pass its output to the real part of complex multiplier 528.

The real part of complex multiplier 528 is configured to accept as its inputs the quadrature output of the interpolator 528 and the quadrature output of the digital local oscillator 530. The real part of a complex multiplication is passed so that the real part of complex multiplier 528 implements only the real output portion of a complex multiplication. The real part of complex multiplier 528 is configured to pass its output to the DAC 532. Still, the invention is not limited in this regard.

In some embodiments of the invention, the digital chaotic sequence and the amplitude-and-time-discrete digital signal are zero intermediate frequency (IF) signals. The digital chaotic sequence is used to amplitude modulate the “known data preamble” and the information symbols via an efficient instantiation of a complex multiplier. The result of this amplitude modulation process is a zero IF signal. Still, the invention is not limited in this regard.

Referring again to FIG. 5, the IF translator and specifically the real part of the complex multiplier 528 are configured to communicate a sampled digital chaotic output signal (i.e., a digital chaotic output signal having an increased sampling rate and non-zero intermediate frequency) to the DAC 532. The DAC 532 is configured to convert a sampled digital chaotic output signal to an analog signal. The DAC 532 is also configured to communicate an analog signal to the anti-image filter 534.

In some applications, it can be desirable to change a sampling rate at the output of the digital complex multiplier 524 only, for example when using an interpolating DAC. An IF translator consisting of an interpolator 526 only can be provided for this purpose.

Referring again to FIG. 5, the anti-image filter 534 is configured to remove spectral images from the analog signal to form a smooth time domain signal. The anti-image filter 534 is also configured to communicate a smooth time domain signal to a RF translator 536. The RF translator 536 is a wide bandwidth analog IF to RF up converter. The RF translator 536 is configured to center a smooth time domain signal at an RF for transmission thereby forming an RF signal. The RF translator 536 is also configured to communicate the RF signal to the power amplifier (not shown). The power amplifier (not shown) is configured to amplify a received RF signal. The power amplifier (not shown) is configured to communicate the amplified RF signal to the antenna element 538 for communication of analog input signal 406 to a receiver 404 (described below in relation to FIGS. 6A and 6B).

It should be understood that the digital generation of the digital chaotic sequence at the transmitter 402 and receiver 404 is kept closely coordinated under the control of a precision real time reference 512 clock. The higher the precision of the clock 512, the closer the synchronization of the chaos generator 300 of the transmitter 402 and the chaos generator (described below in relation to FIGS. 6A and 6B) of the receiver 404 shall be excluding the effects of processing delay differences and channel propagation times. The use of a precision real time reference allows the states of the chaos generators to be synchronized with precision.

Referring again to FIG. 5, the precision real time reference 512 is a stable local oscillator locked to a precision real time reference, such as a GPS clock receiver or a chip scale atomic clock (CSAC). The precision real time reference 512 is configured to supply a high frequency clock to the clocked logic circuits 506 through 532 while being locked to a lower frequency reference clock. The lower frequency reference clock supplies a common reference and a common real time of day reference to prevent a large drift between the states of the chaos generator 300 and the chaos generator (described below in relation to FIGS. 6A and 6B) of the receiver 404 over an extended time interval.

One of ordinary skill in the art will appreciate that the transmitter 402, as shown in FIG. 5, is one exemplary architecture of a transmitter for communications system 400. However, the invention is not limited in this regard and any other transmitter architecture can be used with communications system 400 without limitation. For example, the transmitter 402 can include real first to second intermediate frequency (IF) translation instead of a quadrature first to second intermediate frequency (IF) translation. As another example, other architectures may employ additional chaotic sequence generators to provide a switched chaotic output or to control other aspects of the transmitter 402.

Referring now to FIGS. 6A and 6B, there are provided block diagrams of the receiver 404 of FIG. 4 according to an embodiment of the invention. It should be noted that in conventional analog chaos based coherent communications systems, analog chaos circuits are synchronized by periodically exchanging state information. The exchange of state information requires a substantial amount of additional bandwidth. This additional chaotic state transfer generally makes analog based coherent communications impracticable. The receiver 404 of FIG. 6A is designed to eliminate the drawbacks of conventional analog chaos based coherent communications systems. In this regard it should be appreciated that the receiver 404 is comprised of a digital chaos generator. The receiver 404 includes a tracking loop for synchronizing the digital chaos generator and the digital chaos generator of the transmitter 402. Most significantly, the receiver is configured to synchronize two (2) strings of discrete time chaotic samples (i.e., chaotic sequences) without using a constant or periodic transfer of state update information. A first string of discrete time chaotic samples is generated at the transmitter 402. A second string of discrete time chaotic samples is generated at the receiver 404.

Referring again to FIGS. 6A and 6B, the receiver 404 is comprised of an antenna element 602, a low noise amplifier (LNA) 604, a zonal filter 606, an automatic gain control (AGC) amplifier 608, a radio frequency (RF) to intermediate frequency (IF) conversion device 610, an anti-alias filter 612, and an analog-to-digital (A/D) converter 614. The receiver 404 is also comprised of a quadrature fixed digital local oscillator 660, real multipliers 616, 618, low pass filters 662, 664, a complex multiplier 666 a loop control circuit 620, a quadrature digital local oscillator 622, a correlator 628, multiplexers 646, 648, a channel encoded acquisition data generator (CEADG) 650, digital complex multipliers 624, 652, and a symbol timing recovery circuit 626. The receiver 404 is further comprised of a receiver controller 638, a precision real time reference clock 636, a hard decision device 630, a symbol to bits (S/B) converter 632, and a source decoder 634. The receiver 404 is comprised of a chaos generator 300, a real uniform statistic to quadrature Gaussian statistic mapper (RUQG) 642, and a re-sampling filter 644. Each of the above listed components and circuits 602-318, 622-326, 630-338, 642-352, 660-666, are well known to one of ordinary skill in the art. Thus, these components and circuits will not be described in great detail herein. However, a brief discussion of the receiver 404 architecture is provided to assist a reader in understanding the invention. It should be noted that when the receiver 404 is in both acquisition and tracking modes (described below) the receiver 404 is utilizing a novel architecture/algorithm.

Referring again to FIGS. 6A and 6B, the antenna element 602 is configured to receive the analog input signal 406 generated by the transmitter 402. The antenna element 602 is also configured to communicate the analog input signal 406 to the LNA 604. The LNA 604 is configured to amplify the analog input signal 406 while adding as little noise and distortion as possible. The LNA 604 is also configured to communicate an amplified, analog input signal to the zonal filer 606. Zonal filters are analog filters with slow roll off characteristic but low injection loss used to suppress large interfering signals outside of bands of interest. Zonal filters are well known to one of ordinary skill in the art, and therefore will not be described in great detail herein. It should be appreciated that the zonal filter 606 is configured to communicate a filtered, analog input signal to the automatic gain control (AGC) amplifier 608. An automatic gain control (AGC) amplifier 608 is a controllable gain amplifier used to keep the magnitude of the received signal within normal bounds for the rest of the signal processing chain. Automatic gain control (AGC) amplifiers are well known to one of ordinary skill in the art, and therefore will not be described in great detail herein. It should be appreciated that the automatic gain control (AGC) amplifier 608 is configured to communicate a gain adjusted, analog input signal to the RF to IF conversion device 610.

The RF to IF conversion device 610 is configured to mix the analog input signal to a preferred IF for conversion to a digital signal at the A/D converter 614. The RF to IF conversion device 610 is also configured to communicate a mixed analog input signal to the anti-alias filter 612. The anti-alias filter 612 is configured to restrict a bandwidth of a mixed analog input signal. The anti-alias filter 612 is also configured to communicate a filtered, analog input signal to the A/D converter 614. The A/D converter 614 is configured to convert a received analog input signal to a digital signal. The A/D converter 614 is also configured to communicate a digital input signal to a second IF translator which is comprised of quadrature fixed digital local oscillator 660, the real multipliers 616, 618, the low pass filters 662, 664, the complex multiplier 666, and the programmable quadrature digital local oscillator 622.

The multiplier 616 is configured to receive a digital word as input from the A/D converter 614 and a digital word from the in-phase component of the quadrature fixed digital local oscillator 660. The multiplier 616 multiplies the output of the A/D converter 614 by the in-phase component of the quadrature digital local oscillator 660. The multiplier 616 is also configured to communicate a digital output word. The multiplier 618 is configured to receive a digital word as input from the A/D converter 614 and a digital word from the quadrature-phase component of the quadrature fixed digital local oscillator 660. The multiplier 618 multiplies the output of the A/D converter 614 by the quadrature-phase component of the quadrature fixed digital local oscillator 660. The multiplier 618 is also configured to communicate a digital output word. The quadrature fixed digital local oscillator generates a fixed frequency quadrature signal used to mix the output of the A/D converter 614 to an approximately zero intermediate frequency (IF).

The low pass filter 662 low pass filters the baseband output of multiplier 616. The low pass filter 662 is also configured to communicate a digital output word. The low pass filter 664 low pass filters the baseband output of multiplier 618. The low pass filter 664 is also configured to communicate a digital output word. The output of low pass filters 662, 664 is collectively a quadrature approximate baseband IF signal. The complex multiplier 666 is configured to receive a digital word as input from the low pass filter 662, a digital word as input from the low pass filter 664, a digital word from the in-phase component of the quadrature digital local oscillator 622, and a digital word from the quadrature-phase component of the quadrature digital local oscillator 622. The complex multiplier 666 multiplies the output of the low pass filters 662,664 by the quadrature fine tuning signal of the quadrature digital local oscillator 622. The complex multiplier 666 is also configured to communicate a digital output word.

The quadrature digital local oscillator 622 generates a complex quadrature amplitude-and-time-discrete digital sinusoid at a frequency which shall remove detected frequency and phase offsets in the quadrature baseband IF signal. The quadrature digital local oscillator accepts as its inputs a binary phase control word and a binary frequency control word from the loop control circuit 620. Quadrature digital local oscillators are known to one of ordinary skill in the art, and therefore will not be described in detail herein.

The IF translator is configured to mix the digital input signal to a preferred IF for processing at the correlator 628 and the digital complex multiplier 624. The IF translator is also configured to communicate a digital input signal to the correlator 628 and the digital complex multiplier 624. As will be appreciated by one of ordinary skill in the art, the output of the IF translator can include an in-phase (“I”) data and quadrature phase (“Q”) data. As such, the IF translator can communicate I and Q data to the correlator 628 and the digital complex multiplier 624.

The digital complex multiplier 624 is configured to perform a complex multiplication in the digital domain. In the complex-valued digital multiplier 624, the digital input signal from the IF translator is multiplied by a digital representation of a chaotic sequence. The chaotic sequence is generated in the chaos generator 300, as described with reference to FIG. 3. Thus, the description provided above in relation to FIG. 3 is sufficient for understanding the architecture of chaos generators 300 shown in FIG. 6A. The chaos generator 300 communicates the chaotic sequence to an RUQG 642. In this regard, it should be appreciated that the chaos generator 300 is coupled to the receiver controller 638. The receiver controller 638 is configured to control the chaos generator 300 so that the chaos generator 300 generates a chaotic sequence with the correct initial state when the receiver 404 is in an acquisition mode and a tracking mode.

The RUQG 642 is configured to statistically transform a digital chaotic sequence into a transformed digital chaotic sequence. The transformed digital chaotic sequence can have a characteristic form including combinations of real, complex, or quadrature, being of different word widths, and having different statistical distributions. One such statistical transformation used in the preferred embodiment is a bivariate Gaussian distribution that converts two (2) independent uniformly distributed random variables to a pair of quadrature Gaussian distributed variables. The RUQG 642 is further configured to communicate transformed chaotic sequences to the re-sampling filter 644.

According to the embodiment of the invention, the RUQG 642 statistically transforms a digital chaotic sequence into a quadrature Gaussian form of the digital chaotic sequence. The RUQG 642 communicates the quadrature Gaussian form of the digital chaotic sequence to the re-sampling filter 644. More particularly, the RUQG 642 communicates an in-phase (“I”) data and quadrature phase (“Q”) data to the re-sampling filter 644. Still, the invention is not limited in this regard.

The re-sampling filter 644 is also configured to forward a transformed chaotic sequence to the digital complex multiplier 624. The re-sampling filter 644 is configured as a sample rate change filter for making the chaos sample rate compatible with the received signal sample rate when the receiver 404 is in acquisition mode. The re-sampling filter 644 is also configured to compensate for transmit and receive clock offsets with less than a certain level of distortion when the receiver is in a steady state demodulation mode. In this regard, it should be appreciated that the re-sampling filter 644 is configured to convert a sampling rate of in-phase (“I”) and quadrature-phase (“Q”) data sequences from a first sampling rate to a second sampling rate without changing the spectrum of the data contained in therein. The re-sampling filter 644 is further configured to communicate in-phase (“I”) and quadrature-phase (“Q”) data sequences to the digital complex multipliers 624, 652, and the multiplexers 646, 648.

It should be noted that if a sampled form of a chaotic sequence is thought of as discrete samples of a continuous band limited chaos then the re-sampling filter 644 is effectively tracking the discrete time samples, computing a continuous representation of the chaotic sequence, and resampling the chaotic sequence at the discrete time points required to match the discrete time points sampled by the A/D converter 614. In effect, input values and output values of the re-sampling filter 644 are not exactly the same because the values are samples of the same waveform taken at slightly offset times. However, the values are samples of the same waveform so the values have the same power spectral density.

Referring again to FIG. 6A, the CEADG 650 is configured to generate a modulated acquisition sequence. The CEADG 650 is also configured to communicate a modulated acquisition sequence to the digital complex multiplier 652. The digital complex multiplier 652 is configured to perform a complex multiplication in the digital domain. This complex multiplication includes multiplying a modulated acquisition sequence from the CEADG 650 by a digital representation of a chaotic sequence to yield a reference for a digital input signal. The digital complex multiplier 652 is also configured to communicate the reference signal to the multiplexers 646, 648. The multiplexer 646 is configured to route the quadrature-phase part of a reference signal to the correlator 628. The multiplexer 648 is configured to route the in-phase part of a reference signal to the correlator 628. In this regard, it should be appreciated that the multiplexers 646, 648 are coupled to the receiver controller 638. The receiver controller 638 is configured to control the multiplexers 646, 648 in tandem so that the multiplexers 646, 648 route the reference signal to the correlator 628 while the receiver 404 is in an acquisition mode (described below).

The correlator 628 is configured to correlate a chaotic sequence with a digital input signal. In this regard, it should be understood that, the sense of the real and imaginary components of the correlation is directly related to the values of the real and imaginary components of the symbols of a digital input signal. It should also be understood that in some embodiments, the sense of the real and imaginary components of the correlation is directly related to the values of the real and imaginary components of the PSK symbols of a digital input signal. Thus, when the correlator 628 is in a steady state demodulation mode the output of the correlator 628 is PSK symbol soft decisions. In this regard, it should be appreciated that soft information refers to soft-values (which are represented by soft-decision bits) that comprise information about the bits contained in a sequence. In particular, soft-values are values that represent the probability that a particular bit in a sequence is either a one (1) or a zero (0). For example, a soft-value for a particular bit can indicate that a probability of a bit being a one (1) is p(1)=0.3. Conversely, the same bit can have a probability of being a zero (0) which is p(0)=0.7.

The correlator 628 is also configured to communicate PSK soft decisions to the hard decision device 630 for final symbol decision making or to the source decoder 634 (path not shown). The hard decision device 630 is configured to communicate symbol decisions to the S/B converter 632. The S/B converter 632 is configured to convert symbols to a binary form. The S/B converter 632 is configured to communicate a binary data sequence to the source decoder 634. The source decoder 634 is configured to decode FEC applied at the transmitter and to pass the decoded bit stream to one or more external devices (not shown) utilizing the decoded data.

The correlator 628 is also configured to acquire initial timing information associated with a chaotic sequence, initial timing associated with a data sequence and to track phase and frequency offset information between the chaotic sequence and a digital input signal. The correlator 628 is also configured to track input signal magnitude information between the chaotic sequence and a digital input signal. Acquisition of initial timing information and tracking of input signal magnitude, phase and frequency offset information are all standard functions in digital communication systems. As such, methods for acquiring initial timing information and tracking phase and frequency offset information are well known to one of ordinary skill in the art, and therefore will not be described in detail herein. However, it should be appreciated that any such method can be used without limitation.

Referring again to FIG. 6A, the correlator 628 is configured to communicate the magnitude and phase information as a function of time to the loop control circuit 620. The loop control circuit 620 uses the magnitude and phase information to calculate the deviation of the input signal magnitude from a nominal range, and phase and frequency offset information to synchronize a chaotic sequence with a digital input signal. The loop control circuit 620 is also configured to communicate the phase and frequency offset information to the quadrature digital local oscillator 622 portion of the IF translator and gain deviation compensation information to the automatic gain control (AGC) amplifier 608. The loop control circuit 620 is further configured to communicate a retiming control signal to the re-sampling filter 644 and the chaos generator 300.

It should be understood that the digital generation of the digital chaotic sequence at the transmitter 402 and receiver 404 is kept closely coordinated under the control of a precision real time reference clock 636. The higher the precision of the clock 636, the closer the synchronization of the chaos generator 300 of the transmitter 402 and the chaos generator 300 of the receiver 404 shall be excluding the effects of processing delay differences and channel propagation times. It is the use of digital chaos generators 300 in the transmitter and receiver that allow the states of the chaos generators to be easily controlled with precision, thus allowing coherent communication.

Referring again to FIG. 6A, the precision real time reference clock 636 is a stable local oscillator locked to a precision real time reference, such as a GPS clock receiver or a chip scale atomic clock (CSAC). The precision real time reference clock 636 is configured to supply a high frequency clock to the clocked logic circuits 614, . . . , 666 while being locked to a lower frequency reference clock. The lower frequency reference clock supplies a common reference and a common real time of day reference to prevent a large drift between the states of the chaos generator 300 of the transmitter 402 and the chaos generator 300 of the receiver 404 over an extended time interval while the tracking loops are non active.

The operation of the receiver 404 will now be briefly described with regard to an acquisition mode and a steady state demodulation mode.

Acquisition Mode:

In acquisition mode, the re-sampling filter 644 performs a rational rate change and forwards a transformed chaotic sequence to the digital complex multiplier 652. The CEADG 650 generates a modulated acquisition sequence and forwards the same to the digital complex multiplier 652. The digital complex multiplier 652 performs a complex multiplication in the digital domain. In the digital complex multiplier 652, a modulated acquisition sequence from the CEADG 650 is multiplied by a digital representation of a chaotic sequence to yield a reference for a digital input signal that was generated at the transmitter 402 to facilitate initial acquisition. The chaotic sequence is generated in the chaos generator 300. The digital complex multiplier 652 communicates a reference signal to the multiplexers 646, 648. The multiplexers 646, 648 route the reference signal to the correlator 628. The correlator 628 is transitioned into a search mode. In this search mode, the correlator 628 searches across an uncertainty window to locate a received signal state so that the chaos generator 300 can be set with the time synchronized state vector.

Steady State Demodulation Mode:

In steady state demodulation mode, the correlator 628 tracks the correlation between the received modulated signal and the locally generated chaos close to the nominal correlation peak to generate magnitude and phase information as a function of time. This information is passed to the loop control circuit 620. The loop control circuit 620 applies appropriate algorithmic processing to this information to extract phase offset, frequency offset, and magnitude compensation information. The correlator 628 also passes its output information, based on correlation times terminated by symbol boundaries, to the hard decision block 630. The hard decision block 630 compares the correlation information to pre-determined thresholds to make hard symbol decisions. The loop control circuit 620 monitors the output of the correlator 618. When the loop control circuit 620 detects fixed correlation phase offsets, the phase control of the quadrature digital local oscillator 622 is modified to remove the phase offset. When the loop control circuit 620 detects phase offsets that change as a function of time, it adjusts the re-sampling filter 644 which acts as an incommensurate re-sampler when the receiver 404 is in steady state demodulation mode or the frequency control of the quadrature digital local oscillator 622 is modified to remove frequency or timing offsets. When the correlator's 628 output indicates that the received digital input signal timing has “drifted” more than plus or minus a half (½) of a sample time relative to a locally generated chaotic sequence; the loop control circuit 620: (1) adjusts a correlation window in an appropriate temporal direction by one sample time; (2) advances or retards a state of the local chaos generator 300 by one iteration state; and (3) adjusts the re-sampling filter 644 to compensate for the time discontinuity. This loop control circuit 620 process keeps the chaos generator 300 of the transmitter 402 and the chaos generator 300 of the receiver 404 synchronized to within half (½) of a sample time with the advance/retard control.

More precise temporal synchronization is achieved by the resampling filter 644 which can be implemented as a member of the class of polyphase fractional time delay filters. This class of filters is well known to one of ordinary skill in the art, and therefore will not be described in great detail herein.

In this steady state demodulation mode, the symbol timing recovery circuit 626 communicates a symbol onset timing to the correlator 628 for controlling an initiation of a symbol correlation. The correlator 628 correlates a locally generated chaotic sequence with a received digital input signal during a symbol duration. In this regard, it should be understood that the sense and magnitude of a real and imaginary components of the correlation is directly related to the values of the real and imaginary components of symbols of a digital input signal. Accordingly, the correlator 628 generates symbol soft decisions. The correlator 628 communicates the symbol soft decisions to the hard decision device 630 for final symbol decision making. The hard decision device 630 determines symbols using the symbol soft decisions. Thereafter, the hard decision device 630 communicates the symbols to the S/B converter 632. The S/B converter 632 converts the symbol decisions to a binary form. The S/B converter 632 is configured to communicate a binary data sequence to the source decoder 634. The source decoder 634 is configured to decode the FEC applied at the transmitter 402 and pass the decoded bit stream to one or more external devices (not shown) utilizing the decoded data.

One of ordinary skill in the art will appreciate that the receiver 404, as shown in FIGS. 6A and 6B, is one exemplary architecture of a communications system receiver. However, the invention is not limited in this regard and any other receiver architecture can be used without limitation.

Transmitting Constant Energy Symbols for Reduced BER

As previously described, another aspect of the previous invention is providing a varying symbol length to compensate for variation in symbol energy due to modulation of the carrier using chaotic sequences. In chaotic communications, because the values of the chips used during modulation of each of the symbols vary according to a non-stationary chaotic evolution, in both magnitude and phase, the total energy used during a relatively small fixed interval will also be non-stationary; therefore the energy used to transmit each symbol will vary according to the chaotic sequence. This generally results in a non-stationary energy per symbol to noise power ratio that provides non-optimal BER performance due to nonlinearities in BER curves. Statistical evaluations of the received SNR result in confidence intervals for a chosen BER performance, leading to higher transmission link margins for the entire transmission.

In the various embodiments of the invention, a novel alternative approach is provided for transmitting data symbols spread with a non-stationary spreading sequence, such as a digitally generated chaotic sequence. In particular, the symbols are processed such that each symbol is transmitted with the same symbol energy. Without altering the non-stationary spreading sequence, the only alternative is to dynamically dither the symbol period (effectively dithering the spreading ratio) such that each symbol has the same amount of transmitted energy. Therefore, in the various embodiments of the invention, symbols are transmitted with varying symbol periods to provide a relatively constant symbol energy that results in an improved BER performance for the transmission. In particular, the various embodiments of the invention provide a varying symbol period for improving BER by evaluating the cumulative amount of symbol energy resulting from chips already used for transmitting a current symbol. The transmitter then only terminates transmission of the current symbol if a minimum threshold symbol energy level is met. In other words, the symbol period for each symbol is extended until the cumulative energy provided by the chips for a current symbol reaches at least a configurable minimum value. In the various embodiments of the invention, such a capability can be provided for CCSSS 400 in several ways. In particular, a computation device can be included in transmitter 402 for making a determination that a next symbol should be transmitted. A similar, if not identical, computation device can be included in receiver 404 (or 690) to similarly determine when to transition receiver processing to the next data symbol.

For example, as shown in FIG. 5, a CCSSS providing a relatively constant symbol energy includes a computation device 539 in transmitter 402. The computation device 539 accumulates the energy values of chips being generated by RUQG 520 used for transmitting a current symbol. The computation device 539 is also configured to receive as an input a threshold symbol energy (E_(SYM)). This threshold energy level can be adjusted to provide a desired nominal SNR at the receiver. As one of ordinary skill in the art will recognize, a substantially constant symbol energy will provide optimal BER for all symbols in a transmission, under the assumption of a slow-varying transmission channel. Furthermore, as the amount of symbol energy is increased, as controlled by the re-configurable threshold E_(SYM), the BER is decreased, all other transmission channel parameters held constant.

In operation, the computation device 539 obtains a sum of the energy provided by each of the chaotic chips generated and makes a decision as to whether transmission of the current symbol should be continued or terminated (i.e., transmit a next symbol). If the decision is to terminate transmission of the current symbol, the computation device 539 generates a signal (NEXT SYMBOL) to cause the next symbol to be transmitted. For example, as shown in FIG. 5, the computation device 539 generates a signal for symbol data formatter 506. In such a configuration, symbol data formatter 506 is configured to only provide a next symbol data to be encoded only after a signal is received from computation device 539. However, the invention is not limited to using the symbol data formatter 506. As previously described, the data transfer rate for the data source 502 and the source encoder 504 are also controlled. The signal from computation device 539 can be used to provide a signal for specifying the data rate in such devices or as a data transfer enable control. Alternatively, data source 502 and/data encoder 504 can operate using a non uniform clock signal. In such embodiments, the signal from the computation device 539 can be used as a symbol clock signal for such devices in order to control when bits associated with a next symbol are provided.

In addition to E_(SYM), the computation device 539 can be configured to receive as an input a second threshold value (E_(NOISE)) for use with a selective noise cancellation device (SNCD) within the computation device 539. In such embodiments, the computation device will then include in the total chip sum only chip energy values exceeding a threshold value (E_(NOISE)) to improve SNR. The use of E_(NOISE) and the SNCD will be described below in greater detail.

In another embodiment of the invention, the computation device 539 can be configured (not shown) to communicate a signal (such as NEXT SYMBOL) to the transmitter controller 510 that controls all transmission processes. However, it should be appreciated that any such instantiation of the symbol clock can be used without limitation.

In FIG. 7, there is a block diagram 700 of a computation device 539 according to an embodiment of the invention. As previously described, the computation device 539 can receive the output from RUQG 520. This output is a statistically transformed output of the digital chaotic sequence that forms the chaotic spreading sequence. As a result, each of the chip values of the chaotic spreading sequence are complex numbers with independent Rayleigh distributed magnitudes and uniformly distributed phases. Accordingly, the approximate energy contribution from each of the chips can be computed by multiplying the complex chip values by their conjugates. In another embodiment of the invention, the symbol energy may be calculated directly from the Rayleigh magnitude used in the bivariate uniform to quadrature Gaussian transformation. Both of these processes for calculating the magnitude of a complex number are well understood by those skilled in the art, so will not be discussed here.

Therefore, as shown in FIG. 7, upon receipt of complex chip values “I” and “Q” (collectively “Z”) from RUQG 520, the computation device 539 can first use a conjugate generator 702 to generate the conjugate of chip value “Z”. The chip value “Z” and its conjugate can then be provided to a complex multiplier 704 to determine the energy contribution of the chip. The calculated contribution can then be provided to a resettable summing device 706 configured to accumulate different chip values and sum the accumulated chip values. The sum can then be compared to E_(SYM), where E_(SYM) specifies the minimum symbol energy for a given symbol, as previously described. In the various embodiments, E_(SYM) can be manually provided to the computation device 539 or can be automatically determined based on the configuration of communications system 400. Regardless of how E_(SYM) is provided to the computation device 539, the sum from summing device 706 is periodically compared to E_(SYM) by comparator 708. If the sum from summing device 706 is greater than or equal to E_(SYM), comparator 708 can then generate a signal (NEXT SYMBOL) for the symbol formatter 506. At the same time, comparator 708 generates a CLEAR signal for resetting summing device 706. However, as previously described, the NEXT SYMBOL signal can be provided to any other device within transmitter 402 for signaling transmission of a next symbol. In another embodiment of the invention, the summing device 706 may employ a different thresholding algorithm, many of which are known by those skilled in the art.

In some embodiments, a selective noise cancellation device 710 is provided. In such embodiments, the input E_(NOISE) threshold parameter is provided to the device. If the energy from a chip as calculated in complex multiplier 704 is less than the value E_(NOISE), the value is zeroed before being passed to the summing device 706, effectively cancelling its contribution to the total symbol energy. This mechanism is provided to discount instantaneous symbol energies that are statistically expected to be below the noise floor of a received signal and thus contribute negatively to a symbol decision.

As previously described, transmitter 402 and receiver 404 need to be synchronized in order to properly decode the chaotic communications signal. Although identically configured chaos generators and real time reference systems can be used to synchronize encoding and decoding, a receiver also needs to include a computation device in order to properly determine the beginning and end of symbols in the data signal.

Referring back to FIG. 6A this can be accomplished by the inclusion of computation device 653, respectively. Like computation device 539, computation device 653 from RUQG 642 a locally generated chaos used for decoding a current symbol. The computation device 653 also is also configured to receive as an input a threshold symbol energy (E_(SYM)). This threshold energy level can be identical to that used for computation device 539 and can also be selected to provide a constant SNR in a stationary channel. In operation, the computation device 653 can also obtain a sum of the energy provided by each of the accumulated chips and makes a decision as to whether decoding of the current symbol should be continued or terminated (i.e., decode a next symbol). If the decision is to terminate decoding of the current symbol, the computation device 653 generates a signal (NEXT SYMBOL) to cause the next symbol to be decoded. For example, as shown in FIG. 6A, the computation device 653 generates a signal for symbol timing recovery device 626. In such a configuration, symbol timing recovery device 626 is configured to signal the correlator 628 that a next symbol is to be decoded only after a signal is received from computation device 653. In some embodiments, the computation device 653 can be configured as shown in FIG. 7. However, the invention is not limited in this regard and other configurations for computation device 653 can also be used. For example, in some embodiments, the computation device can generate a signal for the correlator 628. However, as previously described, implementing symbol timing recovery can require additional hardware or configuration of the correlator 628.

One of ordinary skill in the art will appreciate that FIGS. 5, 6A, 6B, and 7 only illustrate exemplary architectures for the various components of the communications system 400 in FIG. 4. However, the invention is not limited in this regard and any other architectures for any of these components can be used without limitation.

In light of the foregoing description of the invention, it should be recognized that the invention can be realized in hardware, software, or a combination of hardware and software. A method of generating a chaotic sequence according to the invention can be realized in a centralized fashion in one processing system, or in a distributed fashion where different elements are spread across several interconnected processing systems. Any kind of computer system, or other apparatus adapted for carrying out the methods described herein, is suited. A typical combination of hardware and software could be a general purpose computer processor, with a computer program that, when being loaded and executed, controls the computer processor such that it carries out the methods described herein. Of course, an application specific integrated circuit (ASIC), and/or a field programmable gate array (FPGA) could also be used to achieve a similar result.

The invention can also be embedded in a computer program product, which comprises all the features enabling the implementation of the methods described herein, and which, when loaded in a computer system, is able to carry out these methods. Computer program or application in the present context means any expression, in any language, code or notation, of a set of instructions intended to cause a system having an information processing capability to perform a particular function either directly or after either or both of the following a) conversion to another language, code or notation; b) reproduction in a different material form. Additionally, the description above is intended by way of example only and is not intended to limit the invention in any way, except as set forth in the following claims.

All of the apparatus, methods and algorithms disclosed and claimed herein can be made and executed without undue experimentation in light of the present disclosure. While the invention has been described in terms of preferred embodiments, it will be apparent to those of ordinary skill in the art that variations may be applied to the apparatus, methods and sequence of steps of the method without departing from the concept, spirit and scope of the invention. More specifically, it will be apparent that certain components may be added to, combined with, or substituted for the components described herein while the same or similar results would be achieved. All such similar substitutes and modifications apparent to one of ordinary skill in the art are deemed to be within the spirit, scope and concept of the invention as defined. 

1. A method for communicating a sequence of information symbols between a transmitter and a receiver using a chaotic sequence spread spectrum signal, the method comprising: generating identical sequences of chaotic chips at said transmitter and said receiver, said sequences of chips synchronized in time; transmitting a sequence of information symbols in signal using said sequence of chips generated at said transmitter, each of said information symbols in said signal having a variable duration of transmission based on a threshold symbol energy value and said chips generated at said transmitter; extracting said sequence of information symbols from said transmitted signal, each of said information symbols in said transmitted signal extracted using said chips generated at said receiver and said threshold symbol energy value; wherein said duration of transmission of each of said information symbols in said signal is a total duration of a selected number of said chips used for transmitting each information symbols, and wherein said number of said chips is selected for each of said information symbols to provide a total chip energy that is greater than or equal to said threshold symbol energy value.
 2. The method of claim 1, wherein said modulating further comprises: selecting a first of said sequence of information data; encoding a current portion of said information data using said selected one of said sequence of information encoding symbols; modulating said current portion of said symbol using a current portion of said identical sequence of chips generated at said transmitter; calculating a total energy of all of said chips used for modulating said selected one of said plurality of information symbols; comparing said total energy for modulating to said threshold symbol energy value; and based on said comparison, ascertaining whether to encode a next portion of said information data using a next of said sequence of information symbols.
 3. The method of claim 2, said ascertaining further comprising: if said total energy for modulating is greater than or equal to said threshold symbol energy value, selecting said next one of said plurality of information symbols, and repeating said encoding, said modulating, said calculating, said comparing, and said ascertaining.
 4. The method of claim 2, said ascertaining further comprising: if said total energy for modulating is less than said threshold symbol energy value associated with said selected one of said plurality of symbols, encoding a next portion of said carrier using said selected one of said plurality of information symbols and repeating said modulating, said calculating, said comparing, and said ascertaining.
 5. The method of claim 2, said calculating further comprising: obtaining conjugate pair product values for each of said chaotic chips in portions of said first of said identical sequences associated with said selected one of said symbols; and obtaining a sum of said conjugate pair product values.
 6. The method of claim 1, wherein said demodulating further comprises: selecting a first portion of said transmitted signal; associating said selected portion of said transmitted signal with one of said sequence of said information symbols; demodulating said selected portion of said transmitted signal using a current portion of said identical sequence of chips generated at said receiver; calculating a total energy of all of said chips used for demodulating said associated one of said sequence of information symbols; comparing said total energy for demodulating to said threshold symbol energy value; based on said comparison, ascertaining whether to associate a next portion of said transmitted signal with said associated one of said sequence of information symbols;
 7. The method of claim 6, said ascertaining further comprising: if said total energy for demodulating is greater than or equal to said selected threshold symbol energy value, decoding said one of said sequence of information symbols; selecting a next portion of said transmitted signal; associating said selected portion of said transmitted signal with a next one of said sequence of information symbols; and repeating said demodulating, said calculating, said comparing, and said ascertaining.
 8. The method of claim 6, said ascertaining further comprising: if said total energy for demodulating is less than said selected threshold symbol energy value, selecting a next portion of said transmitted signal, and repeating said demodulating, said calculating, said comparing, and said ascertaining.
 9. The method of claim 6, said calculating further comprising: obtaining conjugate pair product values for each of said chips used for demodulating said associated one of said sequence of information symbols; and obtaining a sum of said conjugate pair product values.
 10. A system for communicating a chaotic sequence spread spectrum signal, the system comprising: a transmitter for transmitting a sequence of information symbols in a spread spectrum signal using a chaotic sequence of chips generated at said transmitter, each of said information symbols in said signal having a duration of transmission based on a threshold symbol energy value and said chips generated at said transmitter; and a receiver for extracting said sequence of information symbols from said transmitted signal, each of said information symbols in said transmitted signal extracted using a chaotic sequence of chips generated at said receiver and said threshold symbol energy value; wherein said chaotic sequences of chips generated at said transmitter and said receiver are identical and synchronized in time, wherein said duration of transmission of each of said information symbols in said carrier is a total duration of a selected number of said chips used for transmitting each information symbols, and wherein said number of said chips is selected for each of said information symbols to provide a total chip energy that is greater than or equal to said threshold symbol energy value.
 11. The system of claim 10, wherein said transmitter is further configured for: selecting a first of said sequence of information data; encoding a current portion of said information data using said selected one of said sequence of information encoded symbols; modulating said current portion of said symbol using a current portion of said identical sequence of chips generated at said transmitter; calculating a total energy of all of said chips used for modulating said selected one of said plurality of information symbols; comparing said total energy for modulating to said threshold symbol energy value; and based on said comparison, ascertaining whether to encode a next portion of said signal using a next of said sequence of information symbols.
 12. The system of claim 11, wherein said transmitter is further configured during said ascertaining for selecting said next one of said plurality of information symbols, and repeating said encoding, said modulating, said calculating, said comparing, and said ascertaining if said total energy for modulating is greater than or equal to said selected threshold symbol energy value.
 13. The system of claim 12, wherein said transmitter is further configured during said ascertaining for encoding a next portion of said signal using said selected one of said plurality of information symbols and repeating said modulating, said calculating, said comparing, and said ascertaining if said total energy for modulating is less than said selected threshold symbol energy value associated with said selected one of said plurality of symbols.
 14. The system of claim 11, wherein said transmitter is further configured during said calculating for: obtaining conjugate pair product values for each of said chips used for modulating said selected one of said sequence of information symbols; and obtaining a sum of said conjugate pair product values.
 15. The system of claim 10, wherein said receiver is further configured for: selecting a first portion of said transmitted signal; associating said selected portion of said transmitted signal with one of said sequence of said information symbols; demodulating said selected portion of said transmitted signal using a current portion of said identical sequence of chips generated at said receiver; calculating a total energy of all of said chips used for demodulating said associated one of said sequence of information symbols; comparing said total energy for demodulating to said threshold symbol energy value; based on said comparison, ascertaining whether to associate a next portion of said transmitted signal with said associated one of said sequence of information symbols;
 16. The system of claim 15, wherein said receiver is further configured during said ascertaining for: decoding said one of said sequence of information symbols if said total energy for demodulating is greater than or equal to said threshold symbol energy value; selecting a next portion of said transmitted signal; associating said selected portion of said transmitted signal with a next one of said sequence of information symbols; and repeating said demodulating, said calculating, said comparing, and said ascertaining.
 17. The system of claim 16, wherein said receiver is further configured during said ascertaining for: selecting a next portion of said transmitted signal, and repeating said demodulating, said calculating, said comparing, and said ascertaining if said calculated demodulation energy is less than said selected threshold symbol energy value.
 18. The system of claim 16, wherein said receiver is further configured during said calculating to obtain conjugate pair product values for each of said chaotic chips in said second of said identical sequences associated with said one of said symbols and obtain a sum of said conjugate pair product values.
 19. The method of claim 1, wherein said transmitter is further configured to dynamically modify or re-configure the symbol energy threshold.
 20. The system of claim 1, wherein said receiver is further configured to dynamically modify or re-configure the symbol energy threshold. 